After an IC is manufactured, it is tested on an IC tester before it is shipped to a customer. The goal of the testing is to verify that each individual IC was manufactured correctly without defects. At the highest level, testing may be reduced to simply plugging the IC into a host system and determining whether the system appears to be functioning normally while running applications. If the system works, then a determination may be made that the IC is ready to be shipped. However, this type of system-level test does not ensure that the IC is defect free, since the given applications used may exercise only a subset of the IC's functionality. This type of high-level system-based testing also requires a relatively large amount of time.
An alternative to the system test approach is known as functional testing. This type of testing is performed on a general-purpose IC tester (known as Automated Test Equipment, or ATE). This type of testing applies logic 1s and/or 0s to the input pins of the IC in order to stimulate all of the logic gates within the IC, and determines whether each logic gate outputs the correct result by observing the output pins of the IC. The patterns applied to and the results expected from each IC pin are stored in memory on the ATE and exercise the various functional aspects of the IC. If the IC responds correctly to all test stimuli, it is considered to be of shipment quality. However, given the complexity and sequential depth of modern ICs, creating a sufficiently thorough test to be applied via the pins is very difficult, and given the large number of pins on some ICs, the cost of the ATE resources can become prohibitive.
A third alternative to the system testing and functional testing approaches is known as structural testing. Instead of exercising the functional aspects of the IC, this type of testing applies logic 1s and/or 0s internally to stimulate all of the logic gates within the IC, and determines whether each logic gate outputs the correct result, again internally. This internal controllability and observability is obtained by using modified memory elements (flip-flops) inside the IC that are serially connected into a scan chain during test mode. This well-known technique of “scan design” has been in wide use for many years. In “full scan” designs, every internal flip-flop in the IC is made “scannable” by adding a serial access to a predecessor flip-flop and a successor flip-flop on the scan path during test mode. Thus, all the logic gates on the IC are surrounded by scannable flip-flops and become combinationally testable. In order to perform a scan test, data is serially shifted into all of the flip-flops in the scan path while the IC is in test mode, the resulting response of the logic gates to the final scanned-in state stimulus is captured by clocking all the flip-flops one or more times while the IC is in normal mode, and then the newly captured data is serially shifted out of the IC while in test mode. The captured data is analyzed by the ATE as it is shifted out to determine whether the correct results were obtained. The ATE is also responsible for switching the IC between normal and test modes appropriately as well as for providing the clock stimulus.
In order to create a structural test, a software tool uses a simulation model of the IC, which includes the scan flip-flops and all of the combinational logic of the IC. A “fault model” that represents hypothesized defects is superimposed on the simulation model of the IC in order to guide the creation of specific test patterns (also called test vectors) that are intended to expose faulty logic gates. The software tool then generates test patterns for each location in the IC model at which a fault, or defect, could exist. Each test pattern is a set of 1s and 0s that are necessary to excite and propagate the hypothesized fault to an observation point (i.e., a scannable flip-flop), as well as the expected response of a defect-free IC. If an IC responds to such a pattern with data other than that expected, then the hypothesized fault is deduced to be present and the IC is thus determined to be defective and is not shipped. The complete set of test patterns (called a test sequence or a test set) is intended to cover all possible faults in an IC.
The spectrum of all possible faults in an IC is, unfortunately, very broad. While many defects result in permanent logical errors that can be easily detected by scan-based tests, some defects manifest themselves only as increased delays in the IC. Therefore, if the scan test is performed without taking the speed at which the gates should respond into account, such “delay defects” may go undetected. For example, assuming a NOR gate that has a weak pull down transistor, the gate may produce the correct logical value if given enough time, but will not produce the value correctly under the timing specifications for the IC. Therefore, each gate must be checked to determine whether its logical function is correctly performed and whether it is performed in a timely fashion. A pattern that does not take timing into account is called a “static” test, while one that does execute under timing constraints is called a “dynamic” test. A dynamic test for a given logic gate is created by running two test patterns in sequence at full clock speed and determining whether a slow-to-rise (STR) or slow-to-fall (STF) delay fault exists.
For example, if one input of a two-input NOR gate is held at 0 for two clock cycles while the other input changes from a 0 on the first clock cycle to a 1 on the second clock cycle, the output should change from a 1 on the first clock cycle to a 0 on the second clock cycle. If the output does not change from a 1 to a 0 within specified timing margins, a slow-to-fall (STF) fault exists. Similarly, if one input of the NOR gate is held at 0 for two clock cycles while the other input changes from a 1 on the first clock cycle to a 0 on the second clock cycle, the output should change from a 0 on the first clock cycle to a 1 on the second clock cycle. If the output does not change from a 0 to a 1 within specified timing margins, a slow-to-rise (STR) fault exists.
FIG. 1 is a block diagram illustrating a series of scan flip-flops 1, 2, and 3 as well as combinational logic 4 of an IC that incorporates the aforementioned scan design and that can be used for performing static and dynamic tests. Each of the flip-flops 1, 2, and 3 has its data input, D, connected to the output of one of the multiplexers 6, 7, or 8, respectively. When the scan enable signal, SC_EN is low (i.e., not asserted), the data at input 0 of the multiplexers 6, 7, and 8 is captured by the flip-flops 1, 2, and 3, respectively, on the rising edge of the clock, CLK. Therefore, when the scan enable signal SC_EN is not asserted, the IC is functioning in the normal operational manner. The combinational logic 4 will normally receive a plurality of primary input signals 9 originating from the input pins of the IC and will drive a plurality of primary output signals 11 that terminate at the output pins of the IC. The scan chain begins at input S_I 12 and ends at output S_O 13. The flip-flops act as a serial shift register between these two points when the scan enable signal SC_EN is asserted.
The black dots separating flip-flops 2 and 3 are intended to indicate that the scan chain may, and normally does, include many more flip-flops than the three shown in the figure (e.g., 100,000 flip-flops on a contemporary IC is not uncommon). It should be noted that corresponding additional inputs and outputs of the combinational logic 4 are connected to these additional scan flip-flops not shown in FIG. 1 to enable every gate of the combinational logic to be tested using the scan design technique. It should also be noted that several independent scan chains can be used to link groups of flip-flops together instead of one long scan chain as shown.
Given this scan design architecture, several varieties of tests are possible. The first is to establish that the scan chain itself is functional. This is accomplished by asserting the scan enable signal SC_EN (i.e., setting it to logic 1 in this example scan design configuration) and then cycling the clock as many times as there are flip-flops on the scan chain to load the chain, then again that many times to unload the scan chain. The signals at the scan in inputs, S_I, of the multiplexers 6, 7, and 8 are captured by the flip-flops 1, 2, and 3, respectively, on each clock cycle. The first flip-flop on the scan chain captures the S_I input pin 12, and the last flip-flop on the scan chain drives the S_O output pin 13. The scan logic is functioning as one large shift register during this test, with each flip-flop outputting a data value on the rising edge of the clock CLK. During the second half of this test, the expected response of the SO pin 13 should match the values delivered on the S_l pin 12 in the first half of the test.
For example, assuming the scan design logic shown in FIG. 1 comprises 100 flip-flops, 200 clock cycles will be issued to shift a 100-bit test pattern of 1s and/or 0s into the flip-flops and completely out again; SC_EN will be held high over the 200 clock cycles. If all 100 bits appear properly on the S_O pin 13 in order during the second half of the test, then the scan chain is operational.
Once a determination is made that the flip-flops are functioning properly as a scan chain, the combinational logic 4 can be tested. In this case, again assuming 100 flip-flops in the scan design chain, the SC_EN signal will be asserted and 100 clock cycles will be issued to enable a 100-bit test pattern to be shifted into the flip-flops. The final state shifted in at this point in the test is available at the Q-outputs of the flip-flops and corresponds to the stimulus portion of this test pattern. After the combinational logic 4 has settled, the circuit response is available at the 0 inputs of the multiplexers and is captured in the flip-flops by pulsing the clock CLK exactly once with the SC_EN signal held low. Then, by holding the SC_EN signal high and issuing 100 clock cycles, the captured response data is shifted out for analysis. The data shifted out is analyzed by the ATE to determine if the circuit responded to the stimulus properly (against the stored expected response as a reference).
Whether or not this type of test is static or dynamic depends upon the relative timing of the clock signal CLK and the test mode signal SC_EN. A static test would result when there is a pause between the application of the last shift action (the 100th clock pulse with SC_EN high in this example) and the application of the capture event (the single clock pulse with SC_EN low). There are two well-known methods for applying dynamic tests. In the first, known as “last-shift-launch” or “skewed-load” testing, the SC_EN signal must be capable of being switched from high to low in between two at-speed clock pulses (the 100th and 101st, in this example). This would apply two test patterns executed in sequence at speed, with the transitions caused by the last shifted state (from CLK 99 to CLK 100) in test mode being captured by CLK 101 in normal mode (just after SC_EN was lowered at speed). Although this is difficult to do, it has been accomplished in the past by carefully designing the SC_EN signal. Due to the difficulty of making the scan enable signal operate at speed, some designs perform transition testing at reduced speed, with consequent reduction in test quality. Alternatively, the second scheme for implementing dynamic tests, known as “functional justification” or “broadside” delay testing, relaxes this demand on the SC_EN signal, but requires two at-speed clocks during the time when SC_EN is low (i.e. during normal operation). This technique loops through the combinational logic 4, not once, but twice, thereby making the burden on the test pattern generation tools twice as great. For example, after the 100th pulse of the clock CLK with the scan enable signal SC_EN high, SC_EN can be lowered at leisure, then the clock CLK is pulsed twice in succession at full chip speed. The transitions launched on the first of these two clock pulses are captured by the second pulse. The scan enable signal SC_EN is then asserted at any desired time, and then 100 clock pulses are applied to shift out the captured data. It should be noted that the combinational logic 4 was exercised twice during the time when SC_EN was low.
However they are implemented, dynamic tests are intended to detect delay faults. There are two popular delay fault models, namely, transition faults (also known as gate delay faults) and path delay faults. A transition fault models the situation where a single gate in an IC is slow: it will produce the correct logical output, but it will not be able to produce it in a timely fashion. A path delay fault models the situation where several gates are marginally slow, such that any one of them won't adversely affect the ability of the circuit to run at speed, but the combination of all the incremental delays will cause the circuit to be too slow when a path connecting those gates is sensitized.
Practical usage of the transition fault model is impeded by the greedy nature of test generation algorithms, which tend to select the easiest (i.e., the shortest) route into and out of the gate in question, even when there are other (longer) routes that would make the transition test more sensitive to a given delay defect.
Practical usage of the path delay fault model is impeded by the exponential number of paths in a circuit: testing all possible paths in an IC is a mathematical impossibility.
The manner in which scan-based delay fault testing is currently performed has various disadvantages, which will be described with reference to the schematic diagram of FIG. 2. The schematic diagram of FIG. 2 shows a simple example of what might be found in the combinational logic 20 of an IC. The logic 20 comprises, among other logic gates, an AND gate 21, a NOR gate 22, and a NAND gate 23, which are connected such that they form a portion of a logical path connecting circuit nodes B, C, D, and E. Also shown in the figure are two scan flip-flops 24 and 25, as well as a fault location marked with an “X” at the output of NOR gate 22.
Present transition fault test generation algorithms often take advantage of the direct controllability and observability, respectively, afforded by the scan flip-flops 24 and 25 and would launch and capture a rising transition passing through the fault site by using these short paths into and out of the fault site. This choice, unfortunately, reduces the quality of the transition test pattern, since the length of the path through the fault site is not taken into account in determining whether or not a transition fault is detected by a test pattern. For example, assuming a slow-to-rise (STR) fault is located at the “X” on the output of NOR gate 22, a test pattern which produces two consecutive 0s from AND gate 21 at node C on two consecutive clock cycles while launching a falling transition from flip-flop 24 will be able to observe the fault effect at node D via flip-flop 25 on the second clock cycle. Specifically, if the input to flip-flop 25 does not change from 1 to 0 within the second clock cycle, a determination is made that a transition fault exists. If, however, the input to flip-flop 25 does fall from 1 to 0 within the second clock cycle, a determination is made that no defect exists. The problem with this determination is that a delay defect at this fault site could indeed exist but the delay is just too short in duration to affect the particular (short) path Q-D taken to test it. If, alternatively, this fault were tested by holding the output Q of flip-flop 24 at a constant logic 0 while propagating a transition along path A-B-C-D-E-F, or some other relatively long path through the fault site, then the determination that a passing test does in fact indicate the absence of a defect can be made with much more confidence.
Clearly, the transition fault model does not take into account whether a delay defect will cause an IC to fail when the defect is part of a longer path than that taken to test the transition fault at the defect site. The obvious alternative is to generate delay tests by using the path delay fault model on long paths through the circuit. However, the application of the path delay fault model to an entire circuit results in an exponential explosion in the number of paths, rendering this model useless as a general technique for quantifying delay fault coverage. It is generally only applied for a small subset of the longest (critical) paths in a circuit. This results in (optimal) tests for the critical path subset of all chip paths, but does not provide global coverage of delay defects. Rather, only those delay defects on the relatively few selected paths are detected. The path delay fault model also requires that full chip timing analysis be performed to find the worst paths, then discards the bulk of the data produced.
Furthermore, a set of nodes on a path that is functionally unsensitizable may remain untested, even though these nodes could have been tested via another shorter, but truly, sensitizable path. Path identification does not necessarily determine whether a given path is sensitizable. The phrase “sensitizable path”, as that phrase is used herein, is intended to denote any path that can be activated by setting the logic values on gates along the path such that a transition is launched and propagated along the entire path.
The transition fault model, on the other hand, uses a fixed number of fault sites (2 faults per circuit node) and is useful for measuring chip-wide speed coverage. However, the simpler test criteria for the transition fault model allows the incident and departing transition through the target node to propagate through paths in the logic that are relatively short compared to the longest possible path that contains the target node. Any transition test pattern that uses these relatively short paths will be less sensitive in detecting a delay defect than a pattern that uses the longest possible sensitizable path through the target node.
Accordingly, a need exists for a technique that combines the strongest aspects of both of the transition fault model and the path delay fault model in order to provide a test for delay defects that utilizes a bounded set of fault sites and that utilizes the longest sensitizable path through each fault site to perform delay fault testing.